II. STEP BY STEP DESIGN PROCEDURES OF THE DESIRED DUAL BAND FED PRINTED MONOPOLE (DBFPM) ANTENNA

Layout of the desired antenna structure with W _sub × L _sub dimension is indicated in Fig. 1.

Fig. 1. Layout of the desired antenna (dimensions in terms of mm).

The suggested antenna with miniaturized dimension of 15 × 15 mm² has been designed, tooled, and constructed on the FR4 substrate with thickness of 0.8 mm, loss tangent of 0.022 and relative dielectric constant of ε_τ = 4.4. The antenna is fed by a 50 Ω CPW. The main radiator in form of the square shaped patch with the dimension of 7.5 × 7.5 mm² along with a partial CPW ground plane have been made up on the top layer of substrate. The bottom layer includes the modified U-shaped CBP. For the impedance matching, the distance between the radiator and the CPW ground plane is indicated by a gap, which provides the competent control between the lower edge of patch and the CPW ground plane. The optimized gap between the patch and the CPW ground plane is 2.2 mm. To improve the desired antenna performances into two operating bands of WLAN (5.15–5.825 GHz) and X (8–12 GHz), the CBP is modeled and designed in the manner shown in Fig. 1. According to this figure, the conductor-back plane (CBP) is modeled and designed in two steps:

1. Step (1) Build a simple U-shaped CBP (as shown in Fig. 2(a)).

2. Step (2) Insert three strips to step. 1 (so that is shown in Fig. 2(b)).

Fig. 2. Layouts of the CBPs.

The proposed structures based on CBPs are shown in Fig. 2. The U-shaped CBP and the modified U-shaped CBP along with three embedded strips are responsible for making the first band centered at 5.5 GHz and the second band centered at 10 GHz, respectively. The CBP dimensions are regulated and optimized by parametric study to achieve the enhanced impedance matching over the frequency bands. The optimization of the structure is obtained using the HFSS. The optimal parameters of the realized antenna are organized in Table 1.

The other optimized dimensions of the antenna are indicated in Fig. 1.

III. DBFPM ANTENNA DATA

The simulation procedures have been accomplished by applying the electromagnetic software Ansoft HFSS. The antennas were tooled, fabricated and tested at the facility and workshop. The impedance bandwidth of the constructed antenna with proposed novel CBP structure was measured by using an Agilent 8722ES Vector Network Analyzer (VNA). The main problem of the designed antenna is the measuring of its specifications with the VNA, when the coaxial cable is connected to the antenna. If the measurement cable does not decoupled properly, it becomes part of the radiating structure, and thus, changes the input impedance of the antenna. Moreover, the cable distorts the far-field radiation pattern of the antenna by reradiating part of the signal that leaks into the cable. Hence, in this paper a new method for decoupling the cable from the antenna is presented. This method is based on using the ferrite bead.

A common method for minimizing of the cable radiation is employing the ferrite bead on the coaxial cable as close as possible to the feed point of the antenna.

In investigation, a simple monopole element is located at end of the coaxial feed with a ferrite bead positioned on the feed cable. The bead provides a discontinuity in the propagation coefficient for current on the U-shaped CBP, which results in the significant reflection. In the absence of the ferrite bead, the radiation from the cable is directed to below of the U-shaped CBP. When the ferrite bead is positioned on the coaxial cable, the currents in the U-shaped CBP of the coaxial cable are confined to the part above of top of the ferrite bead. It should be noted that, maximum of the radiation efficiency occurs when the bead is λ/4 from the U-shaped CBP. Clearly, the quite minor changes in the position of the ferrite bead can dramatically effect on the radiation efficiency of the antenna. The ferrite bead appears to be a much simpler method of construction for this type of antenna.

To reduction the effect of the cable on the antenna, high-impedance ferrite bead is utilized along the cable near to its connection to absorb the induced power. The measured input impedance curves of the antenna at the operating bands are plotted in Fig. 3.

Fig. 3. Curves of the measured input impedance versus frequency.

In the proposed antenna with introducing the initial U-formed CBP in the manner indicated in Fig. 2(a), the first operating band can be obtained as is shown in Fig. 4. Also, by etching three embedded rectangular strips on the CBP in the manner shown in Fig. 2(b), the second operational band can be achieved as is illustrated in Fig. 4.

Fig. 4. Measured and simulated S ₁₁ of the proposed antenna with dual-band operations.

This figure depicts that there is a good agreement between the simulated and measured results. It is noticed from Fig. 4 that, by embedding the strips in the manner shown in Fig. 2(b), a suitable control on the dual-band operations can be obtained. The effects of the structural element of strips (L _{S 1}) on the antenna's performances are investigated in the Fig. 5. It can be observed that using the structure of Fig. 2(b) and choosing a proper length for the strips (Ls) the antenna's performances have been significantly improved at upper band and the impedance bandwidth of lower band has been enhanced.

Fig. 5. Simulated S ₁₁ for different values of L _{S 1} when W _{S 1} is 0.5 mm.

The simulated and measured peak gain from 4.5 to 6.5 GHz and 8 to 12 GHz for W-LAN and X bands are plot in Fig. 6. In this figure, the antenna gain is proliferated dramatically from about 2 to 4.2 dBi for X-band, whereas the WLAN-band has indicated a gain variation of less than 1 dB between 1 and 1.9 dBi.

Fig. 6. Simulated and measured peak gain for the proposed antenna with dual-band operations.

The measured radiation patterns for four different spot frequencies of 5.2 and 5.8 GHz for WLAN-band and 9 and 11 GHz for X-band in H-plane and E-plane are shown in Fig. 7.

Fig. 7. Radiation patterns of the proposed antenna at the different operational frequencies of 5.2/5.8 and 9/11 GHz for WLAN and X bands, respectively.

Obviously it is understand from Fig. 7 that, this figure approximately exhibits the omni-directional radiation patterns in H-plane and the dipole-like radiation patterns in the E-plane. From an overall view of these radiation patterns, it can know the proposed antenna behaves quite similar to the typical printed monopoles in the operating bands.

To validate of the design processes, the proposed antennas have been compared with conventional antennas and their characteristics are summarized in Table 2.