I. INTRODUCTION
Millimeter-wave (mmWave) frequencies are attractive for a wide range of applications in various fields, including security, safety, health care, industrial quality control, and scientific research.
Frequencies above 100 GHz have lately received increasing attention [Reference Pfeiffer1], based on the rapidly growing need for higher data rates in communication technology. As the available bandwidth is small in classically used low frequency-bands, high data rate communication requires complicated and noise-sensitive modulation schemes. Technology, especially device performance, sets here a natural limit. The alternative is a shift upwards in operational frequency, as above 100 GHz wider bandwidth is available. A set of frequency bands where atmospheric attenuation is low and which are thus well suited for high data rate communication is between 200 and 300 GHz.
Apart from communication, other applications also benefit from the bandwidth available at high-frequencies. On a circuit building block level high power, high bandwidth multiplier chains for signal generation have been demonstrated [Reference Sarmah, Heinemann and Pfeiffer2]. For radar circuits, wider bandwidth improves the range resolution [Reference Statnikov, Sarmah, Grzyb, Malz, Heinemann and Pfeiffer3], leading to more precise measurements. For passive imaging [Reference Dacquay4] enormous bandwidth at high-frequency is of utmost importance, as it is a part of the equation of the noise equivalent temperature difference, which is a measure for detector sensitivity. In this application, cost-effective solutions with silicon technology have the potential to revolutionize the field of security screening.
Traditionally, these high-frequency bands were the domain of III–V semiconductors, due to their superior device characteristics. But with the recent advances in fabrication technology, heterojunction bipolar transistors (HBTs) with a maximum oscillation frequency f max beyond 450 GHz [Reference Heinemann5] have become available. This motivates the development of radio frequency (RF) integrated circuits towards higher frequencies than previously deemed feasible in silicon. The resulting low production costs at high volumes are the major requirement for further growth in emerging consumer markets.
Whether in communication, radar or imaging applications, there is a need to minimize noise in a wireless receiver front-end. Referring to Friis’ formula [Reference Friis6], typically a low noise amplifier (LNA) as a first active component in such a system provides sufficient gain for the incoming signal to reduce the noise impact of subsequent components. This optimizes the signal-to-noise ratio.
On the other hand, not only high gain, but also bandwidth and power consumption are extremely important for the performance of a LNA integrated into a wireless receiver front-end. Therefore all of them have to be optimized, if the LNA is to be used in a real system [Reference Statnikov, Sarmah, Grzyb, Malz, Heinemann and Pfeiffer3].
Despite the advances in silicon germanium technology, gain at higher frequencies is still limited. This creates not only the need for multistage design, but also for optimizing single stage gain. Feedback can provide a compensation for the gain roll-off at elevated frequencies, thus enhancing single stage performance.
To showcase the potential power and reliability of this method, two amplifiers in technologies from different foundries have been designed.
The first presented amplifier is a four-stage small signal amplifier (SSA) in Infineons B11HFC technology [Reference Dilacher, Tiebout, Lachner, Knapp, Aufinger and Sansen7]. It has a high bandwidth and over 20 dB of gain for differential in-system usage at an operational frequency at which, to the authors best knowledge, gain has never been published before in this particular technology [Reference Hamidipour, Jahn, Meister, Aufinger and Stelzer8, Reference Furqan, Ahmed and Stelzer9].
Previously published J-band LNA designs have shown either high gain with high power consumption [Reference Schmalz, Borngräber, Mao, Rücker and Weber10] or high bandwidth [Reference Mao, Schmalz, Borngräber and Scheytt11, Reference Öjefors, Heinemann and Pfeiffer12], but never both in combination. Thus, the second amplifier presented in this work is a four-stage LNA in IHPs SG13G2 technology, that combines high gain with low power consumption, while surpassing the bandwidth limit of the previously published amplifiers with comparable high gain.
This paper is organized as follows: Section II discusses the theory behind using an inductor as a series–series feedback element to boost the gain of a cascode topology. Section III describes the design methodology based on S-parameter analysis, which leads to the circuit architectures from Section IV. Measured results of both circuits are presented in Section V. Finally, Section VI provides a conclusion.
II. SERIES–SERIES FEEDBACK FOR GAIN ENHANCEMENT
In bipolar transistors the optimum collector current density for minimum noise figure, even though increasing with operational frequency, is factors smaller than the necessary current density to reach optimum f max [Reference Voinigescu13]. Thus, biasing the transistor for low-noise operation diminishes single stage gain, but also reduces power consumption, increasing efficiency. To compensate for the low single stage gain, different approaches are possible.
As the unilateral power gain is an inherent figure of merit of a transistor [Reference Gupta14], most approaches are based on reducing the internal feedback of the device, thus providing unilateralization. A common way in CMOS technology is capacitive cross-coupling for neutralization of the pseudo-differential common-source or cascode-structure [Reference Asada, Matsushita, Bunsen, Okada and Matsuzawa15, Reference Fan, Zhang and Sánchez-Sinencio16].
An alternative first step to increase the gain and to extend the operational bandwidth of a mm-wave amplifier is to employ a cascode structure. The usage of a common-base as a load to a common-emitter transistor reduces the Miller capacitance and therefore greatly improves reverse isolation, thus stability and ease of impedance matching.
However, the unilateral power gain is not the absolute maximum achievable gain of a transistor as shown and defined in [Reference Singhakowinta and Boothroyd17]. Contrarily, as previously analyzed for CMOS structures [Reference Hsieh and Lu18], the introduction of feedback at the common-base transistor of the cascode is an option for compensating the gain roll-off at higher frequencies, actually decreasing the reverse isolation and introducing the risk of instability. Simultaneously maintaining unconditional stability thus becomes priority.
Feedback theory based on two port analysis as found in textbooks like [Reference Gray19] describes the underlying working principle of the method applied here through simplification.
Figure 1 breaks down an exemplary cascode consisting of common-emitter transistor Q1, common-base transistor Q2 and feedback inductor L fb into their simplified high-frequency small signal equivalent circuits. Two-port representation is indicated in the figure. Including the additional inductor, the cascode now consists of two cascaded single feedback loops. The inherent Miller-capacitance as part of Y μ is the major feedback element in the common-emitter transistor, best described as shunt-shunt feedback. Modeling the voltage transfer function in terms of a single feedback loop, the feed-forward factor a 1, and the feed-backward factor f 1 are found to be:
$${a_1} = \displaystyle{{{Y_\mu} - {g_m}} \over {({Y_L} + {Y_\mu} + {Y_o})({Y_S} + {Y_\mu} + {Y_\pi} )}},$$
$${f_1} = - {Y_\mu}.$$
Here, Y S is the source admittance and Y L is the combined input admittance of the following common base transistor and the feedback inductor L fb .
Fig. 1. High-frequency small signal equivalent circuit of a cascode. The additional inductor L fb introduces a frequency dependent component to the feedback function, diminishing the phase margin, but also boosting the power gain.
Equation (3) gives the voltage transfer function for the common emitter transistor:
$${A_1} = {Y_S} \cdot \displaystyle{{{a_1}} \over {1 + {a_1}{f_1}}}.$$
At the common-base transistor L fb introduces a feedback path extrinsic to the device. Similar to the case of the common emitter transistor, two-port analysis allows further insight into the working mechanism of the series–series feedback applied here at the common base transistor. As a first step, the Z-parameter-matrices of the common-base transistor and the feedback inductor are added. Extracting the factors a 2 and f 2 for the voltage transfer function A 2 of a single series–series feedback loop as shown in equation (6) results in:
$${a_2} = \displaystyle{{{Z_{fb}}({Z_\pi} {g_m} - 1) + {Z_o}{Z_\pi} {g_m}} \over {({Z_L} + {Z_{fb}} + {Z_o})({Z_S}{Z_\pi} {g_m} - {Z_S} + {Z_{fb}}({Z_\pi} {g_m} - 1) - {Z_\pi} )}},$$
$${f_2} = {Z_{fb}} = j\omega {L_{fb}}.$$
Here, Z S is the output impedance of the preceding common emitter transistor and Z L is the load impedance.
$${A_2} = - {Z_L} \cdot \displaystyle{{{a_2}} \over {1 + {a_2}{f_2}}}.$$
The open loop gain T and the overall voltage transfer function H of the two concatenated single feedback loops then are:
$$T = {A_1} \cdot {a_2} \cdot {f_2},$$
$$H = {A_1} \cdot {A_2} = \displaystyle{{{a_2}} \over {(1/{A_1}) + T}}.$$
As shown in equation (5), both magnitude and phase of a 2 and f 2 change with either increasing frequency or inductance, resulting in a trade-off of phase margin for gain by altering the open loop gain T in such a way that it partly compensates 1/A 1. If now L fb becomes too large, the open loop gain shifts the real part of either the input or output impedance of the circuit to negative values. This is equivalent to S 11 or S 22 rising above unity, which equals instability and potentially oscillation.
Kim et al. [Reference Kim, Yun, Song and Rieh20] investigated the influence of an additional inductance at the base of a common-base transistor amplifier on the maximum stable gain (MSG) and the stability factor k, showing that for inductors of implementable size with regards to accurate modeling, stability can be maintained while increasing the overall gain.
The next section provides further insights into the necessary design methodology based on S-parameter analysis, discussing the benefits and how to avoid the potential risk of instability by employing a frequency dependent feedback network in cascode SSA.
III. DESIGN METHODOLOGY FOR GAIN ENHANCED CASCODES
This section presents the methodology, which was used during the design phase of both amplifiers. It reliably and repetitively produces stable high gain amplifiers at frequencies above 1/3f max of the used technology.
According to the design goal, the basic circuit parameters are chosen. For LNA design, the cascode is biased at the optimal current density for low noise operation J opt . Following this step, the emitter-sizes of the transistors are scaled in such a way that impedance and noise matching can be achieved simultaneously with a simple L-match.
From simulating the optimal feedback inductance value for different transistor sizes it is apparent, that the feedback inductor becomes smaller and therefore harder to model, the bigger the used transistors are. Thus, it is advised to choose the transistor size as small as still suitable to the design goals. Figs 2 and 3 show simulation results for a cascode with transistors in IHP technology with an emitter area of A E = 2 × (0.12 × 0.96) μm2, which is used in the LNA design presented in this work. Here, the inductor is already of parasitic size.
Fig. 2. Influence of the feedback inductor L fb on stability factors k, Δ, and μ as simulated for a cascode in IHP SG13G2 technology at 230 GHz. As the inductor size increases, μ decreases, indicating reduced stability. As k and μ drop below 1, the cascode leaves the area of unconditional stability. The influence of L fb on Δ is negligible.
Fig. 3. Influence of the feedback inductor L fb on the maximum transducer gain G Tmax and the MSG. When k becomes 1, G Tmax equals MSG. Up to this point, it is still possible to conjugately match the cascode at input and output, resulting in improved gain and preserved unconditional circuit stability.
The series–series feedback inductor is introduced at the base of the common-base transistor. As this is a reactive feedback element, the lowest frequency at which the amplifier still has gain becomes important, because at this point the maximum transducer gain is at its highest. Thus, at the low end of the operational bandwidth, the inductor size versus the stability factors k and μ is simulated, see Fig. 2. The continuous decrease of the Edward and Sinsky stability factor μ indicates a reduction in stability as the inductance increases. Simultaneously it is observed, that as Rollet's stability factor k becomes one, the maximum transducer gain G Tmax equals the MSG, see Fig. 3. After this point, the cascode is no longer unconditionally stable, as the load stability circle moves into the Γ L plane. A further consequence of k becoming smaller than one is, that it is no longer possible to achieve a simultaneous conjugated match, thus degrading power transfer and therefore power gain. As is typical for series–series feedback used for gain enhancement, increasing the inductance of L fb also reduces the input and output impedances of the cascode, which changes the size of the output matching elements. This may be particularly critical for power amplifier design, as maximum size transistors are preferred, which especially at frequencies above 160 GHz have low output impedance, even if a cascode topology is utilized [Reference Sarmah, Chevalier and Pfeiffer21].
Following this reasoning and to ensure stable amplifier operation in face of simulation model imperfections, the inductance value has to be chosen conservatively. Rigorous electromagnetic (EM)-modeling becomes necessary at this point, especially if the inductor is small in physical size.
Afterwards, the use of standard formulas for simultaneous conjugated matching as discussed in [Reference Pozar22] provides values for input, output, and interstage matching elements. The inherent loss at high-frequencies of integrated matching elements like transmission lines reduces the achievable gain, but aids the preservation of circuit stability. Thus it is recommended not to compensate for the losses of matching elements by further increasing the feedback.
On the same note, the matching procedure described in [Reference Voinigescu23], while being successfully employed in many LNA designs today, limits the achievable gain due to inductive degeneration. Therefore it is only suited for transistor operation above 1/3f max as long as the used degeneration inductor is small.
If single-stage gain is still too low for the envisioned application, multiple stages can be cascaded. As small transistor sizes with limited current handling capability are used in consecutive stages, linearity problems may arise.
As a last step, every single amplifier stage on its own and all amplifier stages together are simulated not only in the desired frequency-band of operation, but over the whole frequency range, to ensure stability.
IV. CIRCUIT ARCHITECTURE
This section presents the circuit architecture of both amplifiers, that has been the result of following the described design methodology, while highlighting the unique choices for the differing design goals. Last but not least, the passive elements used for on-wafer probing are discussed.
Full-wave EM-simulation results (from HFSS) of the transmission lines and interconnects were included as S-parameter blocks in the simulation during the design phase. In the single amplifier stages, the layout of all elements relevant at higher frequencies is completely symmetrical. This also includes all parasitics, which are introduced through inter-transistor connections or via to the transmission lines and capacitors. These parasitics are incorporated into the simulation post-layout as S-parameter blocks to analyze and correct for their influence on circuit performance.
Both amplifiers consist of four concatenated pseudo-differential cascode stages. A block-level representation of both amplifiers is shown in Fig. 4, including the Marchand baluns and the tuned pads for on-wafer probing.
Fig. 4. A block-level representation of both amplifier architectures. Each amplifier consists of four equal, pseudo-differential stages connected in series, with Marchand baluns and tuned pads on both ends.
Standard resistive biasing is utilized for the cascodes in both cases. The bias network of the common-emitter transistor consists of a current mirror with a resistive connection to the base of the transistor. The resistor has a value of 2 kΩ, which is high enough for the current mirror not to compromise the noise figure of the amplifier. The transistor employed in the current mirror is of half the size of one amplification transistor, thus the resistance at the base of the current mirror transistor has double the value of the resistance at the base of an amplification transistor. The common-base transistor is supplied through a simple voltage divider with a high-frequency short circuit to ground provided by metal-insulator-metal (MIM) capacitors with a cumulative value of 600 fF.
To avoid the influence of via parasitics, a thin strip of metal on the lowest layer of both processes form the small inductor L fb , acting as a reactive feedback element for gain enhancement. During the design period, this metal-strip has been modeled as a shielded transmission line. Its impedance and length have been adjusted to provide high gain, while maintaining circuit stability within the available transistor parameters. Simulation predicts unconditional stability for both amplifiers over the whole frequency range from 1 to 300 GHz.
A) The SSA
Infineons 0.13 μm SiGe BiCMOS technology B11HFC is the basis for the SSA. It features high speed npn-HBTs with an effective emitter width of 0.13 μm, an f T /f max of 250/360 GHz. For these transistors to reach their optimum f max , Infineon recommends the use of a BEBC transistor layout, as employed in this design. Four narrow and two thick copper layers for RF design purposes are available for routing purposes. Additionally, there are MIM capacitors (1.54 fF/μm2) and both TaN- and polysilicon resistors, to cover the whole range of typically needed resistance values.
Each of the four concatenated stages of the SSA consists of a pseudo-differential cascode as presented in Fig. 5. In this technology, the emitter length of the device is adjustable at will. Therefore the transistors use an emitter length of 5 μm, as bigger transistors do not add significant more small signal gain, but increase current consumption considerably. The transistors are biased to operate below the high-current region to further reduce the current consumption, considering their non-minimal size. The ensuing lowered gain is then mitigated through the use of the described series–series feedback method.
Fig. 5. Single stage circuit schematic of the SSA designed in Infineon technology. Input matching is achieved with series transmission lines and shunt capacitors, output matching with a T-match consisting of two transmission lines and a MIM capacitor on each side. An inductor at the base of the common-base transistor provides series–series feedback for gain-enhancement.
The applied feedback inductor L fb has 5 pH of effective inductance, while featuring a post-layout EM-simulated quality factor of 9.
Input impedance matching to a differential 100 Ω reference is done by an L-match consisting of a 18 fF MIM capacitor and a 70 μm transmission line. The output match is provided by a T-match with a 40 μm transmission line at the collector of each common-base transistor, a 37 μm transmission line as a short circuit stub to the supply and a 21 fF DC blocking MIM capacitor.
All four stages draw 65 mA from a 3.3 V supply. The output referred 1 dB compression point was simulated to be at −270 mdBm, using the same SPICE Gummel-Poon transistor model as has been used for all small signal simulations. The SSA uses an area of 580 × 245 μm2; baluns and pads add to that, resulting in an overall area-consumption of 1070 × 270 μm2.
B) The LNA
The LNA is fabricated in IHP's 0.13 μm SiGe BiCMOS technology SG13G2. Besides the 0.13 μm CMOS baseline process with five fine and two thick top patterned aluminum metal layers (2 and 3 μm) for RF applications, this technology is equipped with silicided and unsilicided polysilicon resistors, MIM capacitors (1.5 fF/μm2). The effective emitter width of the HBTs is 120 nm. The f T /f max for this technology is 300/450 GHz.
The presented LNA is designed with four concatenated pseudo-differential cascode stages as shown in Fig. 6. To obtain an optimum noise match to a differential 100 Ω source impedance, the emitter-length of bipolar transistors can be scaled by simply connecting single devices in parallel, while keeping the current density for minimum noise J opt the same Sorin. Following this principle, all four stages employ identical device sizes of A E = 2 × (0.12 × 0.96) μm2. The transistors are used in a BEC configuration, which is the transistor layout recommended by the manufacturer in this technology. The HBTs are biased in a trade-off for low noise versus sufficient gain for suppressing the noise contribution of subsequent stages.
Fig. 6. Single stage circuit schematic of the LNA designed in IHP technology. Input matching is done with a coupled transmission line, output matching with an L-match consisting of one transmission line and a MIM capacitor on each side. R1 and C1 form a filter network for suppressing common-mode oscillations. Similar to the SSA, an inductor at the base of the common-base transistor is used for gain-enhancement.
For this technology, bias point and transistor size, the effective inductance of the feedback inductor L fb is chosen as determined in Section III to be 7 pH. Post-layout EM-simulation shows a Q-factor of 4 for this inductor.
For impedance matching purposes, transmission lines TL1, TL2, and MIM capacitors are used. An unshielded 70 μm long transmission line in the highest metal layer supplies each side of the differential cascode structure. DC-blocking MIM capacitors with a value of 6 fF each complete the output match to differential 100 Ω. Input matching is done with two 20 μm coupled transmission lines.
A filter network composed of R1 and C1 minimizes potential common mode instability due to coupling between the stages through the shared power supply connection.
All four stages of the LNA draw in total 17 mA current from a 4 V supply. Because these transistors are considerably smaller and due to the noise matching procedure biased to use very little current in comparison with the SSA, the output referred 1 dB compression point of this four stage amplifier was simulated to be at −10.9 dBm. The circuit itself is 270 × 280 μm2 in size; each balun adds another 130 × 240 μm2 and the pads increase the break-out length by 95 μm.
C) Passive elements
Input and output Marchand baluns allow the use of single-ended 50 Ω probe contacts. This impedance is then transformed by the baluns to differential 100 Ω. In both technologies, they are implemented with a two-wire coupled transmission line structure in the top thick metalization layer with a 2.4 μm conductor width and a spacing of 2.4 μm. The coupled transmission lines are arranged in a rectangular loop. Their length corresponds to λ/2 at the design frequency, which is 360 μm in this case. A single balun has an EM-simulated loss of 1.5 dB.
For on-waver probing ground-signal-ground pads were added to the single-ended output of the baluns. A short grounded transmission-line stub compensates the pads’ capacitance.
To determine the characteristics of the amplifiers for differential in-system usage, it is necessary to deembed the balun-plus-pad combinations on both sides of an amplifier. For that purpose a breakout structure of two balun-plus-pad combinations were placed back to back in both technology runs.
V. MEASURED RESULTS
Both amplifiers were characterized using ground-signal-ground (GSG) waveguide probes for the 220–325 GHz band by GGB (GGB Industries Inc.), OML (Oleson Microwave Labs Inc.) mmWave VNA extender modules and an Agilent E8361A network analyzer. For on-wafer characterization, on-chip baluns have been added at both the input and the output. As extracted from the breakout structures mentioned in the previous section, the measured average loss due to the pads and baluns is around 2.5 dB per side from 200–280 GHz in both technologies. Figure 7 presents micrographs of both amplifiers.
Fig. 7. Top chip micrograph: The SSA uses 580 × 245 μm2 of area; baluns and pads add to that, resulting in an overall area-consumption of 1070 × 270 μm2. Bottom chip micrograph: The area of the LNA without pads and baluns, e.g. for differential in-system usage, is 280 × 270 μm2. The whole circuit breakout area is 730 × 360 μm2.
A) The SSA
The SSA in Infineon's B11HFC technology has 19.5 dB of gain at 212 GHz with a measured 3 dB bandwidth of 21 GHz, see Fig. 8. If the pad and balun losses are deembedded, the gain is 5 dB higher and comes out at 24.5 dB. Measurement equipment limits the accurate determination of the bandwidth on the lower side, as the probes and OML modules are only specified between 220–320 GHz. Unconditional stability is attested through the Edward & Sinsky stability factor μ, which is above unity in the whole measured frequency range, as can be found in Fig. 9. Equally, the reverse isolation is around −30 dB in the frequency band of interest, as shown in Fig. 10.
Fig. 8. Measured and simulated small signal gain of the amplifier in Infineon technology in comparison. The measured and simulated data both include the baluns and tuned RF pads at the input and output.
Fig. 9. Measured stability factor μ for the four-stage Infineon amplifier, indicates unconditional stability. For noise reduction purposes, the measured data has been averaged. The simulated μ of the whole four-stage SSA including baluns is equally above unity.
Fig. 10. Measured reverse isolation and reflection of the four-stage Infineon SSA including baluns and pads.
B) The LNA
Figure 11 presents the measured and simulated small signal gain S 21 in comparison for the LNA implemented in the IHP SG13G2 process. They are in good agreement, which validates the quality of the EM-modeling used for simulation of the matching elements. The four-stage amplifier has 22.5 dB maximum gain at 233 GHz. The 3 dB-bandwidth is 10 GHz. Correcting for pad and balun losses, the LNA has a net gain of 27.5 dB for differential in-system usage, e.g. to drive a mixer directly from an on-chip differential antenna. The input and output return losses S 11 and S 22 are plotted in Fig. 12, together with the reverse isolation S 12. The Edward & Sinsky stability factor, calculated from the measured S-parameters, indicates unconditional stability over the whole measured frequency range, as shown in Fig. 13.
Fig. 11. Measured and simulated small signal gain of the LNA in IHP technology in comparison. The measured data includes the baluns and tuned RF pads at the input and output, the simulated data includes balun losses only.
Fig. 12. Measured reverse isolation and reflection of the four-stage LNA including baluns and pads.
Fig. 13. Measured stability factor μ for the four-stage LNA, indicates unconditional stability. For noise reduction purposes, the measured data has been averaged. The simulated μ of the whole four-stage LNA including baluns is equally above unity.
Because of limitations in measurement equipment, specifically the unavailability of a high-frequency noise source and a fundamental J-band mixer, the actual noise figure has not been measured. Instead Fig. 14 presents the simulated noise figure over frequency. Due to the noise impedance match to the source impedance at the design frequency of 230 GHz, it comes out at 12.5 dB. At the bottom of Fig. 7 a micrograph of the 280 × 270 μm2 small 233 GHz LNA can be found.
Fig. 14. Simulated noise figure of the four-stage LNA. The notch at the design frequency confirms the matching of the noise impedance to the source impedance.
VI. CONCLUSION
Two amplifiers in differing SiGe BiCMOS technologies have been presented. Both rely on the deliberate use of feedback for gain-enhancement to achieve high gain above 200 GHz. Employing a 0.13 μm SiGe BiCMOS process currently under development for the SSA and a 0.13 μm SiGe BiCMOS process available through MPW for the LNA, this paper demonstrates the feasibility of further exploration of high mmWave frequencies, potentially up to the THz region. Table 1 summarizes the results of all J-band LNA implementations the authors are aware of. To the authors’ best knowledge, the gain and bandwidth achieved with the SSA is the highest ever reported in that particular technology. For the LNA, both 22.5 dB as a measured breakout structure and the deembedded 27.5 dB gain for differential in-system usage are the highest gain ever reported in any SiGe technology in this frequency range.
Table 1. Comparison of mmWave LNAs.
CB, common base; NF, noise figure; BW, bandwidth.
ACKNOWLEDGEMENTS
This work was partly funded by the European Commission within the project DOTSEVEN (no. 316755). The authors would like to thank Neelanjan Sarmah, University of Wuppertal, Germany, for his help during the measurements and Philipp Hillger and Felix Lelchuk, also from the University of Wuppertal, for fruitful discussions.
Stefan Malz received his Master of Science (Electrical Engineering) from the University of Wuppertal in 2012. Since then, he is with the Institute for High-Frequency and Communication Technology. His main research interests are design and optimization of highly integrated millimeter- and sub-mm wave circuits.
Bernd Heinemann received his diploma degree in Physics from the Humboldt University of Berlin, Germany, in 1984, and the Dr.-Ing. degree in Electrical Engineering from the Technische Universität Berlin, Germany, in 1997. He joined the IHP in 1984. Since 1993 he is a member of a team working on the exploration and technological implementation of SiGe HBTs.
Rudolf Lachner received his Diploma and Ph.D. degrees in Physics from Technical University Munich, Germany, in 1978 and 1984, respectively. As a Senior Principal in the field of RF technologies his main research interests are currently in pushing the limits of SiGe technology into the Terahertz region and paving the way to broad usage of Si based mm-wave technology in new safety and communication applications.
Ullrich R. Pfeiffer received his Ph.D. in 1999. In 2007 he received a European Young Investigator Award. Dr., Pfeiffer was a co-recipient of the 2004, 2006, and 2012 Award for Outstanding Paper at the ISSCC, a co-recipient of the 2006 IBM Pat Goldberg Memorial Best Paper Award, the 2008 EuMIC, 2009 RFIC, and 2010 EuMC Best Paper Award.
