I. INTRODUCTION
Since U.S. Federal Communication Commission (FCC) released ultra-wideband (UWB) technology for commercial communication applications with unlicensed [1] to the public, the study on UWB technology has never been interrupted. UWB bandpass filter (BPF) plays an important role in UWB technology, such as medical imaging system, internet of things. It is, hence, highly demanded for various UWB communication systems. Unfortunately, some undesired wireless radio frequency signals, such as WLAN at 5.8 GHz, C-band satellite signals at 5.975–6.475 GHz, 6.725–7.025 GHz, may interfere with UWB systems. Therefore, UWB BPF with single or multiple notch bands is necessary.
Generally speaking, there are two main kinds of approaches for notch design in UWB filter. One kind of the methods introduces extra notch circuits in the UWB filter design. The other kind of methods utilizes wave cancellation theory to generate transmission zeroes in the passband of UWB filters. Many different extra notch circuits have been proposed [Reference Li, Su, Zhang, Wang and Liang2–Reference Wong and Zhu16], which include additional single/multiple mode notch resonator [Reference Zhu and Chu4–Reference Sekar and Entesari9], defected ground structure [Reference Li, Su, Zhang, Wang and Liang2, Reference Chen, Wu and Liang3, Reference Karimi, Khaminhamedani and Siahkamari10, Reference Chen, Jiao, Xie and Zhang11], embedded open stubs [Reference Xu, Kang, Miao and Wu12], and metamaterial resonators [Reference Wei, Wu, Shi and Chen13, Reference Ali and Hu14]. In this method, UWB filters with single/multiple notch band are designed generally in two steps. A UWB filter is designed and then circuits notched at the desired band are added to the UWB filter, which reduces the efficiency of the design and increases the overall size of the filter compared with the original UWB filter. The wave cancellation theory based methods use asymmetric coupled feed lines [Reference Kim and Chang15, Reference Wong and Zhu16] or two parallel filters to provide two paths for wave cancellation [Reference Mohammadi, Valizade, Nourinia and Rezaei17, Reference Gholipoor, Honarvar and Virdee18]. The overall size of the wave cancellation based UWB filters are reduced dramatically compared with the extra notch circuit based UWB filters. However, the wide tunable notch band of the wave cancellation based UWB filter is difficult to achieve since the phase difference of the two travelling signals should be (2n + 1)π when they arrive a port. Besides the above approaches, UWB filters using intrinsic zeroes to achieve notch band is also practical [Reference Chen, Zhang and Peng19, Reference Wang, Tam, Ho, Kang and Wu20].
In this paper, we propose a new quad-mode resonator (QMR) for designing a UWB BPF with a notch band. The notch band is centered at a frequency of 6.25 GHz to eliminate the C-band satellite signals. It is shown that a notch band is generated due to the intrinsic zero, which is produced by the QMR. The main advantages of the proposed filter is that it not only simplifies the notch band design procedure by utilizing the intrinsic zero but also provides abundant poles/zeros to improve passband/stopband performance, and such benefits as circumventing multilayer operation and via holes. In addition, wide tunable notch band form 5– 9.3 GHz can be achieved by properly changing the dimensions of the loaded cross-shaped resonator. Odd and even-mode analysis method is applied to analyze the QMR. Finally, a UWB notch band BPF is designed, fabricated, and measured. It shows that the measurement results of the UWB notch band BPF have high attenuation in notch frequency and low return loss in passbands, which are in agreement with the full-wave simulation results.
II. CHARACTERISTICS OF THE QMR
A) Resonance properties
As shown in Fig. 1(a), the proposed QMR consists of T-shaped step impedance resonator (TSSIR) (denoted by (l 1, w 1) and (l 2, w 2)) loaded with cross-shaped open stubs (denoted by (l 3, w 3), (l 4, w 4), and (l 5, w 5)), and directly coupled by interdigital feeding structures. It is convenient to adopt the odd and even-mode analysis method to analyze the QMR, due to the fact of symmetry topology of the structure. If the odd-mode is excited, the symmetrical plane T–T′ can be considered as an electric wall, where the stub end is shorted. If the even-mode is excited, the symmetrical plane T–T′ can be considered as a magnetic wall, where the stub end is opened. The odd and even-mode equivalent circuits are shown in Figs 1(b) and 1(c), respectively. Using the transmission line theory, we have
$$Y_{in - odd} = - jY_2\cot (\theta _2),$$
$$Y_{in - even} = Y_2\displaystyle{{Y_{in{\rm 3}} + j\tan \theta _2Y_2} \over {Y_2 + j\tan \theta _2Y_{in3}}},$$
$$Y_{in3} = Y_{in2} + j{\rm tan}\theta _{\rm 1}Y_{\rm 1},$$
$$Y_{in2} = Y_{\rm 3}\displaystyle{{Y_{in{\rm 1}}{\rm +} j{\rm tan}\theta _{\rm 3}Y_{\rm 3}} \over {Y_3 + j{\rm tan}\theta _{\rm 3}Y_{in{\rm 1}}}},$$
$$Y_{in{\rm 1}}= j\left( {{\rm tan}\theta _{\rm 4}Y_{\rm 4} + {\rm tan}\theta _{\rm 5}Y_{\rm 5}} \right),$$
where, Y n (n = 1, 2, …, 5) and θ n (n = 1, 2, …, 5) denote the characteristic admittance and electrical length, respectively. From the resonance conditions of Y in-even = 0 and Y in-odd = 0, we have
$$\eqalign{& {Y_2\left( \matrix{k_1k_5\tan \theta _5\tan \theta _3\tan \theta _1 + k_5\tan \theta _5\tan \theta _3\tan \theta _2 \hfill \cr \quad + k_4\tan \theta _4\tan \theta _3\tan \theta _2 + k_1k_4\tan \theta _4\tan \theta _3\tan \theta _1 \hfill} \right.}\cr & \left. {-k_3k_5\tan \theta _5 + k_3k_4\tan \theta _4 + k_3^2 \tan \theta _3 + k_3\tan \theta _2 + k_1k_3\tan \theta _1} \right)/ \cr & \quad \left( {k_1k_5} \right.\tan \theta _1\tan \theta _2\tan \theta _3\tan \theta _5 \cr & \qquad + k_1k_4\tan \theta _1\tan \theta _2\tan \theta _3\tan \theta _4 - k_5\tan \theta _3\tan \theta _5 \cr & \qquad - k_4\tan \theta _3\tan \theta _4 + k_3 - k_3k_5\tan \theta _2\tan \theta _5 \cr &\qquad - k_3k_4\tan \theta _2\tan \theta _4\cr &\qquad - k_3^2 \tan \theta _2\tan \theta _3 - k_1k_3\tan \theta _1\tan \theta _2 \big) = 0,}$$
$$\cot \left( {\theta _2} \right) = 0,$$
$$Z_{in} = - \displaystyle{{Y_{\rm 4}{\rm tan}\theta _{\rm 3}{\rm tan}\theta _{\rm 4} + Y_{\rm 5}{\rm tan}\theta _{\rm 5}{\rm tan}\theta _{\rm 3} - Y_{\rm 3}} \over {{\rm 2}jY_{\rm 3}{\rm (}Y_{\rm 3}{\rm tan}\theta _{\rm 3} + Y_{\rm 4}{\rm tan}\theta _{\rm 4} + Y_{\rm 5}{\rm tan}\theta _{\rm 5}{\rm )}}}.$$
For simplicity, the center frequency is fixed at f 0 = 6 GHz, 2Y 1 = Y 2 = ⋯ = 2Y 5 = 1/140, thus, k n = 1/2 (n = 1, 3, 5), k 4 = 1, where, k n = Y n /Y 2 (n = 1, 3, 4, 5). Figure 2 depicts the variation of f z1/f 0, f e1/f 0, f o1/f 0, f e2/f 0, f notch/f 0, f e3/f 0, and f z2/f 0 versus l 5, l 4, l 3, and l 1 based on equations (6) and (7) and Z in = 0. It can be observed in Fig. 2(a) that f e2/f 0, f z1/f 0, f notch/f 0, and f e1/f 0 decrease as l 5 increases. However l 5 has almost no effect on f o1/f 0, f e3/f 0, and f z2/f 0. As shown in Fig. 2(b), l 4 has slightly effect on f e1/f 0, f z1/f 0, and no effect on f o1/f 0, and f z2/f 0. In addition, f notch/f 0, f e2/f 0, and f e3/f 0 will decrease with the increase of l 4. The variation of l 3 mainly impacts on f z1/f 0, f e1/f 0, f notch/f 0, and f e3/f 0, while f o1/f 0, f e2/f 0 and f z2/f 0 keep almost unchanged. The variation of l 1 has almost no effect on f e1/f 0, f o1/f 0, f e2/f 0, f z1/f 0, and f notch/f 0, while f e3/f 0 and f z2/f 0, decrease as l 1 is getting larger. Here, four resonant modes named f e1, f e2, f e3, and f o1 have been excited, which can be divided by intrinsic zero into two groups, i.e. the first three modes and the fourth mode. With the interdigital coupled structure feeding, the first three modes will form the UWB lower passband, as shown in Fig. 3(a), the fourth pole together with another two poles generated by interdigital coupling structure form the UWB upper passband. Figure 3(b) gives the comparison of |S 21| between TSSIR and QMR. The effect of interdigital structure on notch frequency in terms of g 2, fl, and fw are given in Fig. 4. It can be seen g 2, fl, and fw mainly impact on the bandwidth of the notch band and attenuation at notch frequency, while the notch frequency keeps almost unchanged.
Fig. 1. (a) Layout of the proposed QMR structure, (b) even-mode equivalent circuit, (c) odd-mode equivalent circuit.
Fig. 2. Variation of f z1/f 0, f e1/f 0, f o1/f 0, f e2/f 0, f notch/f 0, f e3/f 0 and f z2/f 0 versus (a) l 5, (b) l 4, (c) l 1, and (d) l 3.
Fig. 3. (a) Resonance characteristics of the proposed QMR, (b) comparison of |S 21| between TSSIR and QMR.
Fig. 4. The effect of interdigital structure on notch frequency with the variation of (a) g 2, (b) fl, (c) fw.
B) Generation of notch band
As shown in Fig. 5(a), the frequency response of |S 21| with varied θ 4 is investigated, as well as the input impedance Z in (Z in = 1/Y in ) of the loaded cross-shaped open stubs as indicated in Fig. 1(a). From Fig. 5(a) we can see that the transmission zero f z1 has almost no change, while f notch shifts down when the θ 4 changes from 46° to 67°. It is worth to note that Z in = 0 wherever there is a transmission zero. Considering the case that Y 3 = Y 4 = Y 5, the θ 5 and θ 3 are fixed in equation (8), we can derive the flowing results from Z in = 0.
$$f_{notch} = \displaystyle{c \over {2\pi \sqrt {\varepsilon _{re}} l_4}}(\alpha _1 - \alpha _2).$$
Here, α 1 = 1/tan(θ 3), α 2 = tan(θ 5), c, and ε re denote the velocity of light in free space and effective dielectric constant of the stub with w 4, respectively. It can be seen from (9) that the notch frequency shifts down when the physical length l 4 is enlarged. In addition, wide tunable notch band form 5–9.3 GHz can be achieved by properly changing the dimensions of the loaded cross-shaped resonator, as shown in Fig. 5(b). Therefore, the notch band can be generated by setting Z in = 0 at the specified frequency, which can short out the T-shaped step impedance resonator at f notch and finally generate the notch band.
Fig. 5. (a) The frequency responses of |S 21| and Z in with varied θ 4, (b) |S 21| of the UWB with a notch band for different dimensions of the loaded cross-shaped resonator. The direction of the arrow means that the dimensions of the loaded cross-shaped resonator are increasing.
For further demonstration, the current density distribution of the QMR at the notch frequency of 6.6 GHz is shown in Fig. 6. It can be seen that the current is highly covered on the stub l 3, l 4, l 5, and left side of stub l 1. It can be observed that the stubs of l 3, l 4, and l 5 can absorb the electromagnetic wave signal at 6.6 GHz, which blocks the signals transmission from one port to another port.
Fig. 6. Current density distribution of the QMR at 6.6 GHz.
III. IMPLEMENTATION OF UWB NOTCH BAND BPF
In order to provide co-existence of UWB and other systems, an example of UWB BPF with one fixed notch band at 6.25 GHz is designed and fabricated on RO4003 with ε r = 3.55 and h = 0.508 mm, which is shown in Fig. 7(a). The dimensions of the filter are given as follows (all in millimeter):
Fig. 7. (a) Photograph of UWB BPFs with a notch band and (b) simulated and measured results of the UWB notch filter.
l 1 = 4.1, l 2 = 15.4, l 3 = 3.6, l 4 = 10, l 5 = 9.6, fl = 8, g 1 = 0.4, g 2 = 0.1, w 1 = 0.1, w 2 = 0.1, w 3 = 0.1, w 4 = 0.2, w 5 = 0.6, fw = 0.1. The overall size of the filter is 15.4 × 17.6 mm2 (excluding feeding lines). Figure 7(b) plots the simulation and measurement results of the filter, from which we can see that they are in good agreement. It shows that the return loss of simulation results is better than 22 and 20 dB in lower and upper passband, respectively. It can also be observed from Fig. 7(b) that the 3-dB bandwidth of the measurement results ranges from 3.14 to 10 GHz, and the attenuation of the upper stop band is larger than 20-dB until 15 GHz. The measured return loss is better than 14.5 and 11.2 dB in the lower and upper passband, respectively. The minimum insertion loss (including the SubMiniature version A (SMA) connector insertion loss) is as low as 0.3 and 0.69 dB in lower and upper passband, respectively. The four poles generated by the QMR can be found at 3.5, 4.7, 5.07, and 8.8 GHz while the other two poles produced by the interdigital coupled structure are located at 7.2 and 9.65 GHz. Two zeros introduced by the QMR locate at 2.6 and 11.7 GHz, which sharpen the passband edges. The notch band centers at 6.25 GHz with attenuation of 22 dB. Besides, extremely narrow notch band can be found in the measurement results, which its 10-dB fractional bandwidth (FBW) is around 3.84%. Comparison between some published UWB notch band BPFs and this work is listed in Table 1 in terms of poles, return loss, notch frequency/attenuation, transmission zeros in lower and upper stopband, 3-dB FBW, shorted circuit element required, using extra circuit to create notch band and overall size.
Table 1. Comparison with some proposed UWB BPF with notch band.
λ 0 is the free space wavelength at 6.85 GHz.
IV. CONCLUSION
A UWB filter with a notch band centered at 6.25 GHz has been designed and fabricated based on a new QMR without using any other circuits for notch generating. The characteristics of the proposed QMR have been analyzed using odd and even-mode analysis method. It shows that the notch band can be generated by one of three intrinsic zeros produced by the QMR itself, and other two zeros are used to sharpen the passband edges. A good agreement between measured and simulated results verified the proposed UWB notch band BPF.
ACKNOWLEDGEMENT
This work was supported in part by the Macao Science and Technology Development Fund under Projects 032/2010/A2 and 068/2013/A2.
Yang Xiong received the B.Eng. degree from Shandong University of Technology, Zibo, China, in 2013, and is currently working toward the Ph.D. degree at Nankai University, Tianjin, China. His main research interests include microwave circuits design, high temperature superconducting terahertz devices, and terahertz signals detecting.
Cheng Teng received the B.Sc. and M.Sc. degrees in Electrical and Computer Engineering from University of Macau, Macau, China in 2013 and 2016, respectively. From November 2013 to June 2016, he was a Research Assistant with the Faculty of Science and Technology, Wireless Communication Laboratory, University of Macau. His main research interests include UWB bandpass filters and tunable bandpass filters.
Ming He received the M.Eng. and Ph.D. degrees in Electrical Engineering from Nankai University, Tianjin, China, in 2002 and 2008, respectively. Since 1997, he has been with the Department of Electronics, Nankai University, Tianjin, China, where he became a Professor in 2016. From 2004 to 2009, he held several visiting research appointments in Juelich Research Center and Karlsruhe University, Germany. His research interests are in the fields of the applications of high temperature superconductors, the generation and detection of THz signals, and microwave communications.
Kam-Weng Tam (S'91–M'01–SM'05) received the BSc and joint Ph.D. degrees in Electrical and Electronics Engineering from the University of Macau, Taipa, Macao, China, and the University of Macau and Instituto Superior Técnico (IST), Technical University of Lisbon, Lisbon, Portugal, in 1993 and 2000, respectively. Since 1996, he has been with the University of Macau, where he is currently a Professor, the Associate Dean (Research and Graduate Studies) of the Faculty of Science and Technology. He has co-authored over 100 journal and conference papers. His research interests have concerned multifunctional microwave circuits, RFID, UWB for material analysis. Dr. Tam supervised two IEEE MTT-S Undergraduate Scholarship recipients in 2002 and 2003. He was the founder of the IEEE Macau AP/MTT Joint Chapter in 2010 and was chair in 2011–2012. He was a member of the organizing committees of 21 international and local conferences including co-chair of APMC2008.
Xian Yu received the B.Eng. degree from Yanshan University, Qinhuangdao, China, in 2013, and the M.E. degree from Nankai University, Tianjin, China, in 2016. Her main research interests are microwave filters and antennas design.
Wai-Wa Choi (M'00–SM'10) was born in Macau in 1970. He received the B.Sc., M.Sc., and Ph.D. degrees in Electrical and Electronics Engineering from the University of Macau, Macau SAR, China, in 1993, 1997, and 2008, respectively. From 1993 to 1995, he was with Institute of Systems and Computer Engineering (INESC), Lisbon, Portugal, as a Research Assistant. Since 1995, he has been with the University of Macau, Macau, China, where he is currently an Associate Professor in the Department of Electrical of Computer Engineering. He has authored or coauthored over 40 internationally refereed journal and conference papers. His research interests are in the areas of microwave active and passive circuits, smart antennas, radar, and RFID systems. He has been the Chair of IEEE Macau AP/MTT Joint Chapter since 2015.
Hai Hua Chen (S'06-M'16) received the M.Eng. degree in Electrical Engineering from Nankai University, Tianjin, China, in 2002 and the Ph.D. degree in Electrical and Electronic Engineering from University of Hong Kong in 2006. From 2007 to 2010, she joined the Communication Systems Group, Darmstadt University of Technology, Darmstadt, Germany, where she was a Postdoctoral Fellow. Beginning from September 2010, she has been with the College of Information Technical Science, Nankai University, Tianjin, China, where she currently holds an Associate Professor position. Her research interests include microwave communications, distributed beamforming and resource allocation in wireless relay networks, broadband sensor arrays, and statistical array signal processing.
