I. INTRODUCTION
Tunable filters are in demand because of the potential to simplify the radio frequency (RF)-frontend architecture of wideband receivers and to realize software-defined radio (SDR) systems [Reference Gomez-Garcia, Sanchez-Soriano, Tam and Xue1–Reference El-Tanani and Rebeiz4]. Several tunable single-ended (unbalanced) bandpass filters (BPFs) have been presented in different technologies. In general, the tuning range is limited to one octave or less because of the limitation of the capacitance ratio (CR) used in the tuning elements. To go beyond this limitation, band-transition approaches were proposed in [Reference Cho and Rebeiz5, Reference Lin and Rais-Zadeh6]. However, the previous approaches were limited to only two-band (low-band and high-band) transition and could not continuously cover a wide tuning frequency range, e.g. the entire ultra-high-frequency (UHF) band. Besides band-transition approaches, the stacked three-filter bank approach was reported in [Reference Sun, Kaneda, Baeyens, Itoh and Chen7] and realized continuous tuning from 100 to 620 MHz. However, the input and output (I/O) of the three tunable filters are separated, which is not efficient for compact implementation.
Moreover, the balun BPF with center-frequency tunability, which converts an input unbalanced single-ended signal into its balanced ones while maintaining the high-frequency selectivity of a tunable BPF, is a promising candidate for a miniature SDR module [Reference Zhou, Tang, Chen and Bao8–Reference Li and Xue10]. Although several works on a balun BPF with a fixed center frequency on various technologies, e.g. PCBs (printed circuit boards) [Reference Mao and Chueh11–Reference Wang, Huang, Zhu and Wu14], LTCCs (low-temperature co-fired ceramics) [Reference Tamura, Ishizaki and Höft15], IPDs (integrated passive devices) [Reference Chen, Huang and Horng16], and GaAs and CMOS processes are reported, only a few works on a tunable balun BPF are reported. These tunable ranges are limited because of the limitation of the CR.
In this paper, we present a three-band switchable and tunable BPF with a nested two-filter bank structure to achieve a continuous wide-frequency tuning range, i.e. nearly the entire UHF band (400 MHz to 3 GHz), with a common I/O port for convenient use without increasing the footprint. In addition, we also present a two-band switchable and tunable balun BPF using the same technique for the proposed band-switchable and tunable BPF.
The proposed nested tunable BPF and band-switchable tunable balun BPF are implemented with coupled microstrip line switchable λ/4 or λ/2 resonators and RF-micro-electromechanical systems (MEMS) capacitors and achieve significant improvements with regard to frequency tuning range.
The rest of this paper is organized as follows: first, single-ended and balun tunable BPF designs using the tunable band-switching technique are described. Next, their practical implementations are described, and finally, their measurement results are presented.
II. TUNABLE BPF DESIGN
A) Three-band switchable and tunable single-ended design
The proposed tunable filter consists of two switchable RF-MEMS-tuned resonators (internal coupled line resonators and external coupled line resonators with switches) as shown in Fig. 1(a). Using this architecture, a three-band passband transition for coarse-frequency tuning is achieved with the combination of switch states as shown in Fig. 1(b). The details are described as follows.
Fig. 1. Basic elements of proposed nested tunable filters. (a) Architecture and (b) three-band switchable filter concept.
When the RF switches in external resonators (switch 1a, 1b) and those in internal resonators (switch 2a, 2b) are turned on, these internal resonators act as the quarter-wavelength resonators for the lowest frequency band (f 1), and the passband switches to f 1. While the RF switches in both external and internal resonators are turned off, the internal resonators act as the half-wavelength resonators for the middle-frequency band (f 2), which is nearly 2f 1, and the passband switches to f 2. Finally, when the RF switches in only internal resonators (switch 2) are turned on, the internal resonators act as the half-wavelength resonators for the highest frequency band, which is nearly 3f 1, and the passband switches to f 3. Thus, a three-band passband transition for coarse-frequency tuning is achieved. In addition, fine-frequency tuning at each band is achieved with tunable capacitor C v , e.g. the RF-MEMS and varactor diode.
B) Continuously wide tunable design with nested structure
To expand the tunable frequency range and achieve a continuous wide-tuning range with a common I/O port without increasing the footprint, we propose a three-band switchable and tunable BPF with a nested filter bank structure as shown in Fig. 2(a). In this design, the high-band switchable BPF shown in Fig. 1(a) is nested in a low-band switchable BPF so that the size of the nested filter bank is almost the same as that of the low-band tunable filter.
Fig. 2. Proposed continuously tunable filter. (a) Nested architecture and (b) five-band switchable and continuously tunable filter concept.
Although maximally a six-band transition, i.e. f L1, f L2, and f L3 for the low-band BPF and f H1, f H2, and f H3 for the high-band BPF, can be achieved with the proposed nested filter bank, achieving continuous tuning range of nearly the entire UHF band is possible with just five bands, i.e. f L1, f L2, f L3, f H1, and f H2. Figure 2(b) shows such an example of the spectrum allocation of the prototype filter. Thus, we adapted a five-band switchable design to avoid design complexity.
C) Two-band switchable and tunable balun BPF design
In a general RF front-end, the balun is widely used for I/O ports of differential circuits, and the functions of the balun and tunable single-ended BPF are individually implemented as a cascaded circuit as depicted in Fig. 3(a). In this architecture, all implemented circuit sizes tend to be large. Thus, to reduce circuit size, several works on a balun BPF with a fixed center frequency using various technologies have been reported. Although for SDR systems, the function of center-frequency tunability as depicted in Fig. 3(b) is also required to realize frequency agile devices, only a few works on a tunable balun BPF are reported [Reference Zhou, Tang, Chen and Bao8–Reference Li and Xue10].
Fig. 3. Architectures of tunable balun filters. (a) Cascaded filter/line/balun circuit architecture and (b) incorporated tunable balun filter architecture.
In this article, we propose a two-band switchable and tunable balun BPF using the same technique for the proposed three-band switchable and tunable BPF.
The architecture of the proposed balun BPF is shown in Fig. 4(a). The tunable balun BPF consists of a pair of switchable RF-MEMS-tuned resonators with switch 3. Compared with the three-band switchable BPF, switch 1 attached to the external resonator is eliminated, and an additional port (port 3) is added as an unbalanced port on the side of the external resonator. Using this architecture, a two-band passband transition for coarse-frequency tuning is achieved by changing the switch states as shown in Fig. 4(b). The details are described as follows.
Fig. 4. Basic elements of proposed two-band switchable and tunable balun filters. (a) Architecture and (b) two-band switchable filter concept.
When the RF switches in internal resonators (switch 3a, 3b) are turned off, the internal resonators act as the half-wavelength for the lower frequency band (f B1), and the passband switches to f B1. While the RF switches in the internal resonators are turned on, the internal resonators act as the 3/4-wavelength for the higher frequency band (f B2), which is nearly 1.5 f B1, and the passband switches to f B2. These band sates correspond to the band states of middle band (f 2) and the highest band (f 3) of three-band switchable single-ended design. Thus, the two-band passband transition for coarse-frequency tuning is achieved as shown in Fig. 4(b). In addition, fine-frequency tuning at each band is achieved with tunable capacitor C v , the same as the proposed tunable single-ended BPF.
III. IMPLEMENTATION OF THE PROPOSED BPF
A) Design 1: 0.4–3-GHz tunable nested BPF
The filter bank was built on RT/Duroid 4050B with ε r = 3.5 and h = 0.76 mm with an overall size of 26.0 × 36.0 mm2 (0.09 × 0.06 λ g 2, λ g : guided wave-length) not including tunable capacitor controllers. Because of the high-density assembly requirements, the low- and high-band tunable BPFs are formed in a nested structure. To accomplish a common I/O port for the filter bank, external coupling lines are formed face-to-face. The structure parameters for the low-band switchable BPF shown in Fig. 5(a) are: L 1 = 10.0 mm, L 2 = 10.3 mm, L 3 = 12.8 mm, L 4 = 9.8 mm, L 5 = 1.0 mm, L 6 = 7.0 mm, W 1 = 1.7 mm, W 2 = 1.0 mm, W 3 = 2.5 mm, W 4 = 0.2 mm, W 5 = 0.2 mm, g 1 = 0.15 mm, and g 2 = 0.15 mm. In addition, those for the high-band switchable BPF shown in Fig. 5 (b) are: L 1 = 4.9 mm, L 2 = 7.4 mm, L 3 = 6.2 mm, L 4 = 4.0 mm, W 1 = 1.3 mm, W 2 = 0.5 mm, W 3 = 0.2 mm, W 4 = 0.2 mm, g 1 = 0.15 mm, and g 2 = 0.15 mm.
Fig. 5. Filter structure parameters. (a) Low-band switchable BPF and (b) high-band switchable BPF.
Figure 6 shows the filter layout showing the position of the RF-MEMS integrated circuit (IC), RF switches, and direct current (DC) bias networks. Coupling lines are connected with ultra-small surface mount coaxial connectors (HIROSE U.FL-R-SMT-1) through an SPDT switch (Infineon BGS12AL7-4). To achieve good performance on insertion loss and tunability, high-Q and large capacitor tuning range RF-MEMS digital capacitor ICs (WiSpry WS1050), which integrate a three-capacitor bank in an IC, are used. They also include a serial interface (SPI or MIPI® RFFE) and charge pump for providing the 40-V operating voltage. The tuning range of each capacitor bank is 0.5–5.6 pF. In this prototype, two-capacitor banks of the RF-MEMS capacitor IC are used as tuning capacitances for each tunable BPF; thus, two RF-MEMS capacitor ICs are used in total. RF-MEMS capacitor ICs are controlled via WiSpry's interface control boards, which apply a supply voltage (1~3.3 V) and SPI/MIPI® RFFE controlling signal to the ICs. In this work, to prove the switchable band transition concept, resonators remain as open stubs, and via halls are formed near the end of open stubs. The 0-Ω resistors are placed to realize an ideal switch-on condition by connecting between open stubs and via halls. AVX 0604 capacitor lumped components are used for bypass capacitors.
Fig. 6. Filter layout showing positions of RF-MEMS IC, RF switches, and DC bias networks.
Figure 7 (a) and (b) show the simulated S-parameters for different switch conditions and values of C v. When the open stubs in both external and internal resonators and via halls are connected, that is, they are considered as being in a switch on-state, a lowest frequency passband (f L1, f H1) is achieved. While open stubs in both external and internal resonators remain, that is, they are considered as being in a switch off-state, a middle-frequency passband (f L2, f H2) is achieved. Finally, when open stubs in only internal resonators are connected to via halls, the highest frequency passband (f L3, f H3) is achieved. In this design, f L2 and f L3 are designed to be continuously tuned with the use of tunable capacitors, and f H1 is designed to complement the gap between f L1 and f L2. In addition, to expand the tunable frequency range, f L3 and f H2 are also designed to be continuously tuned.
Fig. 7. Simulated S-parameters for different capacitor values in (a) insertion loss and (b) return loss.
In summary, five passbands of f L1, f H1, f L2, f L3, and f H2 are designed to be continuously tuned with the use of the combination of RF-MEMS-tuned switchable resonators and the nested structure. Thus, the simulated nested tunable BPF achieves a 0.4–3-GHz continuous wide tuning range.
B) Design 2: 1.1–1.7-GHz tunable balun BPF
The balun filter was also built on RT/Duroid 4050B with ε r = 3.5 and h = 0.76 mm with an overall size of 37.0 × 39.0 mm2 not including tunable capacitor controllers. The structure parameters of the switchable balun BPF shown in Fig. 8 are: L 1 = 10.0 mm, L 2 = 20.0 mm, L 3 = 24.0 mm, L 4 = 15.0 mm, L 5 = 3.3 mm, L 6 = 17.5 mm, L 7 = 1.0 mm, W 1 = 2.0 mm, W 2 = 0.3 mm, W 3 = 2.0 mm, W 4 = 2.0 mm, W 5 = 0.4 mm, W 6 = 1.0 mm, g 1 = 0.15 mm, and g 2 = 0.15 mm.
Fig. 8. Balun filter structure parameters.
Figure 9 shows the balun filter layout showing the position of the RF-MEMS IC, RF switches, and DC bias networks. External coupling lines of balanced ports are formed in a hair-pin resonator and are connected with a coupling capacitor (C m ) and ultra-small surface mount coaxial connectors (HIROSE U.FL-R-SMT-1). An unbalanced port (port 1) is connected with SMA coaxial connectors. To achieve good performance on both the passband and broadband tunable range, the RF-MEMS digital capacitor IC (WiSpry WS1050) is used. In this prototype, two-capacitor banks of the RF-MEMS capacitor IC are used as tuning capacitances for the tunable balun BPF. RF-MEMS capacitor ICs are controlled via WiSpry's interface control boards.
Fig. 9. Filter layout showing positions of RF-MEMS IC and DC bias networks.
In this work, to prove the switchable band transition concept, resonators remain as open stubs and via halls are formed near the end of open stubs. The 0-Ω resistors are placed to realize an ideal switch-on condition by connecting between open stubs and via halls. AVX 0604 capacitor lumped components are used for bypass capacitors.
Figure 10(a) and (b) show the simulated mixed-mode S-parameters ((a) differential mode response: S ds , (b) common mode response: S cs ) for different switch conditions and values of C v , respectively. The corresponding simulated return losses (S 11) are shown in Fig. 11. When the open stubs in internal resonators are connected to via halls, that is, when switches are considered as being in a switch on-state, a higher frequency passband (f B2) is achieved. While open stubs in internal resonators remain, that is, when switches are considered as being in a switch off-state, a lower band (f B1) is achieved. In this design, f B1 and f B2 are designed to be continuously tuned with the use of tunable capacitors. The simulated center frequency of the passband continuously tunes from 1.1 to 1.75 GHz with simulated insertion loss from 1.4 to 5.7 dB and simulated return loss from 9 to 15.6 dB not including the loss of micro coaxial cables as shown in Fig 10(a) and (b), and the corresponding 3-dB band widths are 75–280 MHz. In addition, the simulated common-mode response is maximally suppressed below 30 dB within the passband.
Fig. 10. Simulated mixed-mode-S-parameters for different capacitor values in (a) S ds and (b) S cs .
Fig. 11. Simulated return loss for different capacitor values.
Figure 12(a) and (b) show the simulated phase difference between balanced ports and magnitude imbalance. Phase imbalance and magnitude imbalance in f B1 from 1.1 to 1.43 GHz are better than 5o and 0.60 dB, respectively. In addition, the simulated phase imbalance and magnitude imbalance in all passbands including the higher passband (f B2) from 1.1 to 1.75 GHz are better than 20o and 1 dB, respectively.
Fig. 12. Simulated (a) phase difference and (b) magnitude imbalance.
The common-mode rejection ratio (CMRR) is usually used to evaluate the balance performance of a balun [Reference Eisenstadt, Stengel and Thompson17]. It is defined as the ratio between the differential-mode (S ds ) and common-mode transmission-coefficient (S cs ) magnitude
$$CMRR = \left \vert {\displaystyle{{S_{ds}} \over {S_{cs}}}} \right \vert = \left \vert {\displaystyle{{\left( {S_{21} - S_{31}} \right)/\sqrt 2} \over {\left( {S_{21} + S_{31}} \right)/\sqrt 2}}} \right \vert. $$
For a balun (splitter-phase shift equal to 180o), CMRR as a function of amplitude and phase-imbalance terms becomes
$$CMRR = \left \vert {\displaystyle{{S_{ds}} \over {S_{cs}}}} \right \vert = \left \vert {\displaystyle{{1 + \left( {\displaystyle{{1 - \Delta /2} \over {1 + \Delta /2}}} \right)e^{ - j\theta}} \over {1 - \left( {\displaystyle{{1 - \Delta /2} \over {1 + \Delta /2}}} \right)e^{ - j\theta}}}} \right \vert, $$
where θ and Δ represent the phase imbalance and magnitude imbalance, respectively.
The simulated CMRR for this design shown in Fig. 13 exceeds 25 dB in the lower passband from 1.1 to 1.4 GHz and exceeds 15 dB in all passbands including the higher passband (f B2) from 1.1 to 1.75 GHz.
Fig. 13. Simulated CMRR.
IV. MEASUREMENT RESULTS
A) Design 1: 0.4–3-GHz tunable nested BPF
Figure 14 shows a picture of the fabricated tunable BPF with the WiSpry's SPI/RFFE interface control board. The interface control board is connected to a PC via a mini-USB cable to adjust capacitor value. The U.FL to SMA micro coaxial cables are connected to I/O ports of the fabricated tunable BPF.
Fig. 14. Fabricated tunable filter with WiSpry's interface control board.
Figure 15(a)–(c) show the measured S-parameters for different switch conditions and values of C V . Although the measured passband frequencies are slightly lower than the simulated ones especially in f L3 and f H2, the filter continuously tunes from 0.4 to 3.0 GHz with a measured insertion loss from 3.2 to 6.8 dB and a measured return loss from 12.7 to 38.7 dB not including the loss of micro coaxial cables as shown in Figure 15(a)–(c).
Fig. 15. Measured S-parameters for different capacitor values in (a) insertion loss, (b) insertion loss around lower bands (whole work band), and (c) return loss.
The input third-order intercept point (IIP3) is also measured at the different center-passband frequencies. Figure 16(a) characterizes the IIP3 at 600 MHz. The IIP3 is measured with two tones located at 599.5 and 600.5 MHz, respectively. Thus, the IM3 is located at 598.5 MHz. The measurement result exhibits an IIP3 of 37 dBm as shown in Fig. 16(a). Generally, an IIP3 of a tunable filter is mainly degraded due to the non-linearity of the tunable capacitors. Thanks to the high linearity of RF-MEMS capacitors, a higher IIP3 than that of the varactor diode tuned filters is obtained. The extracted IIP3 at a 1-MHz offset versus different center frequencies is also shown in Fig. 16(b). The IIP3 ranges from 17.5 to 37 dBm as the passband frequency varies from 400 MHz to 3 GHz.
Fig. 16. (a) Measured output power versus input power at 600 MHz and (b) measured IIP3 versus tuned frequency.
B) Design 2: 1.1–1.7-GHz tunable balun BPF
Figure 17 shows a picture of the fabricated tunable balun BPF board. The SPI/MIPI interface control board is connected to a PC via a mini-USB cable to adjust the capacitor value. The U.FL to SMA micro coaxial cables are connected to the balanced output ports of the fabricated tunable balun BPF.
Fig. 17. Fabricated balun tunable filter.
Figure. 18(a) and (b), and Fig. 19 show the measured mixed-mode S-parameters ((a) differential mode response: S ds , (b) common mode response: S cs ) and S 11 for different switch conditions and values of C v, respectively. Although the measured passband frequencies are slightly lower than the simulated ones especially in the higher passband (f B2), the filter continuously tunes from 1.1 to 1.68 GHz with measured insertion loss from 3.1 to 9.1 dB and measured return loss from 9 to 40 dB not including the loss of the micro coaxial cables as shown in Figs 18 and 19, and the corresponding 3-dB band widths are 100–250 MHz.
Fig. 18. Measured S-parameters for different capacitor values in (a) Sds and (b) Scs.
Fig. 19. Measured return loss for different capacitor values.
Figure 20(a) and (b) show the measured phase difference and magnitude imbalance. The measured phase imbalance and magnitude imbalance in the lower passband (f B1) from 1.1 to 1.32 GHz is better than 12o and 0.50 dB, respectively. In addition, the measured phase imbalance and magnitude imbalance in all passbands including the higher passband (f B2) from 1.1 to 1.68 GHz is better than 28o and 1.5 dB, respectively. As shown in Fig. 21, the corresponding measured CMRR for this design exceeds 20 dB in the lower passband from 1.1 to 1.32 GHz and exceeds 11.2 dB in all passbands including the higher passband (f B2) from 1.1 to 1.68 GHz.
Fig. 20. Measured (a) phase difference and (b) magnitude imbalance.
Fig. 21. Measured CMRR.
V. PERFORMANCE SUMMARY
A) Design 1: 0.4–3-GHz tunable nested BPF
Table 1 compares the performance parameters of the tunable filter in this work with reported filters in the UHF band, and Fig. 22 shows the figure of merit (FOM) versus the tuning ratio (T.R.). Here, the T.R. and FOM are defined by the following equations, respectively [Reference Pleskachev and Vendik18].
$$T.R. = \displaystyle{{f_{{\rm max}}} \over {f_{{\rm min}}}}, FOM = \displaystyle{{f_{{\rm max}} - f_{{\rm min}}} \over {\sqrt {\Delta f_1 \Delta f_2}}} \displaystyle{1 \over {\sqrt {IL_1 IL_2}}}, $$
where f max and f min are the highest and the lowest tuning frequency, and IL 1,2 are the insertion loss, and Δf 1,2 are the 3-dB bandwidth at f max and f min, respectively.
Fig. 22. FOM of tunable BPFs in UHF band.
Table 1. Comparison of reported tunable filter performance in UHF band
BW, bandwidth; I.L., insertion loss.
As shown in Fig. 22, the presented tunable planar filter of this work offers the widest tuning range, which is two times that of Ref. [Reference Lin and Rais-Zadeh6] with a two-band switchable resonator concept, and the highest FOM containing a moderate filter size, and a competitive high linear performance.
B) Design 2: 1.1–1.7-GHz tunable balun BPF
Figure 23 compares the FOM versus the T.R. of the tunable balun filter in this work with reported tunable balun filters and a tunable differential BPF. As shown in Fig. 23, without the proposed band-switching technique, the simulated and measured tuning ratios are <1.3. Whereas, with the use of the proposed technique, the simulated and measured T.R. improved to 1.6. Corresponding to that, the simulated and measured FOM also improved to 1.7 and 0.8, respectively. Although the T.R. and FOM are lower than that of Ref. [Reference Zhou, Tang, Chen and Bao8], the insertion loss of the balanced ports of Ref. [Reference Zhou, Tang, Chen and Bao8] are asymmetric especially under the passband, and the reduction under the passband is not enough, whereas the stopband under the passband of this work is always more than 30 dB as shown in Fig. 18(a).
Fig. 23. FOM of tunable balun BPFs and a differential BPF.
Furthermore, to compare the performances of the implemented balun BPFs with those of pervious designs, the values of the in-band CMRR associated with amplitude and phase imbalances, as defined in equation (2), are illustrated in Fig. 24 with some CMRR contour lines of 15–30 dB. Although in higher band mode CMRR degrades up to 11.2 dB due to the asymmetric topology, the maximum measured CMRR in the lower band mode is better than 25 dB, which is a typical CMRR for the conventional fixed balun BPFs, and the lowest CMRR contains 20 dB. Moreover, because the total CMRR of circuit system is determined by entire circuit topology [Reference Yang, Caloz and Wu19], the higher band mode of the balun BPF can be also useful with the proper circuit system design.
Fig. 24. Comparison of measured CMRR for various works.
IV. CONCLUSION
This paper proposed and verified a wide tunable range UHF BPF for SDR transceivers with the use of switchable RF-MEMS-tuned resonators. The tunable BPFs and the tunable balun BPF were realized with RF-switches for coarse-tuning and RF-MEMS tunable capacitors for fine-tuning. To achieve a continuous tuning range without increasing the filter size, low-band and high-band tunable BPFs were constructed in a nested filter bank structure. The fabricated nested tunable BPF achieved a 0.4–3-GHz continuous tuning range, and the fabricated tunable balun BPF achieved 1.1–1.7-GHz continuous tuning range.
ACKNOWLEDGEMENTS
This research was performed under the research contract of “Research and Development on control schemes for utilizations of multiple mobile communication networks”, for the Ministry of Internal Affairs and Communications, Japan.
Keiichi Motoi received his B.S. and M.S. in Physics from Keio University, Yokohama, Japan, in 2008 and 2010, respectively. He joined NEC Corporation in 2010 where he has been engaged in the research and development of RF modules and RFIC circuits for wireless communication. His current interests include high-efficiency power amplifier architecture and IC/module implementation for mobile base stations and multi-mode/multi-band transceiver ICs for software-defined-radio systems. He is a member of the Institute of Electronics, Information, and Communication Engineers (IEICE), Japan.
Naoki Oshima received his B.E. and M.E. in Electrical Engineering from the Doshisha University, Japan, in 2008. After receiving his M.E., he joined NEC in 2008, where his research interests are RF circuit design for wireless communications, especially ultra-wideband and millimeter-wave applications. He is a member of the Institute of Electronics, Information, and Communication Engineers (IEICE), Japan.
Masaki Kitsunezuka (M'12) received his B.S. in Physics from Keio University, Tokyo, Japan, in 2003 and M.S. in Physics from the University of Tokyo, Japan, in 2005. He joined NEC Corporation in 2005 and has been engaged in research and development for reconfigurable RF CMOS circuits for software-defined and cognitive radio systems. From 2012 to 2013, he was a visiting industrial fellow at the University of California, Berkeley, CA, USA. He is currently a principal researcher in the System Platform Research Laboratories, NEC Corporation. He is a member of the IEEE and of the Institute of Electronics, Information, and Communication Engineers (IEICE), Japan.
Kazuaki Kunihiro received his B.S. and M.S. in applied physics from the Tokyo Institute of Technology, Tokyo, Japan, in 1988 and 1990, respectively, and D.E. in Quantum Engineering from Nagoya University, Nagoya, Japan, in 2004. In 1990, he joined NEC Corporation, Kawasaki, Japan, where he has been engaged in device simulation, modeling, and circuit design of GaAs FETs, InGaP/GaAs HBTs, and GaN FETs for wireless communications. From 1995 to 1996, he was a visiting researcher at the Technical University of Berlin, Berlin, Germany, where he studied non-linear modeling of III–V compound semiconductor devices. His current interests include high-efficiency power amplifier architecture, such as envelope tracking and digital transmitters and their IC/module implementation for mobile base stations, and multi-mode/multi-band transceiver ICs for software-defined-radio systems. Dr. Kunihiro is a member of the Institute of Electronics, Information, and Communication Engineers (IEICE), Japan.
