I. INTRODUCTION
In recent years, compact size dual-band antennas with simple structures are extremely needed for different wireless communication applications. Several designs for dual-band antennas have been introduced and studied [Reference Quan, Li, Cui and Tentzeris1–Reference Quan, Li, Jin and Tentzeris13]. In [Reference Quan, Li, Cui and Tentzeris1], an analysis and design of a compact dual-band directional antenna for 2.4/5-GHz wireless access point and RFID reader application is proposed. The antenna realizes a high gain however it occupies a large space of 43 ×26 × 12 mm3 with a large ground plane of 100 × 60 mm2. In [Reference Zhu, Guo and Wu2], a novel dual-band antenna for wireless communication applications is fabricated. The antenna is fed by a coaxial probe, which increases its size and increases also the complexity level of its fabrication comparing with coplanar waveguide (CPW) feed technique adopted in our designs. In [Reference Zhu, Guo and Wu3], a compact dual-band antenna for wireless body-area network applications is fabricated. The antenna suffers from narrow bandwidths of only 1.2 and 2.0% at the lower and upper bands, respectively. Tunable dual-band antennas for 0.7–1.1-GHz and 1.7–2.3-GHz carrier aggregation systems are presented in [Reference Avser and Rebeiz4]. The antennas solve the drawback of narrow bandwidths by using varactor diodes. However, the efficiencies are found to be poor mainly due to the RF power dissipated by the varactors. On the other hand, designing dual-band antennas with large frequency ratios, which operate in different frequency bands may be required for some applications such as satellite communications [Reference Smith, Gothelf, Kim and Breinbjerg5] and ISM bands [Reference Feng and Leung6]. In this paper, we will introduce the design of two dual-band antennas based on the mechanism of magnetic dipoles obtained by shorting one edge of the radiating patches to the ground plane [Reference He, Gong and Gao7]. The first designed antenna has a large frequency ratio of 2.58 compared with that one obtained in [Reference Chen, Wang and Wu8–Reference Li, Yang and Nie11]. In order to enhance the impedance bandwidth of the upper band, a dual-band antenna is designed based on the modification of the RF feed line structure. The technique avoids the use of lumped components, which leads to an acceptable maximum measured gain of 6.1 dBi with a maximum measured efficiency of 74%. A measured bandwidth value of 9.23% is obtained at the central frequency 6.5 GHz, which is much higher compared with the values obtained in [Reference Zhu, Guo and Wu3, Reference Avser and Rebeiz4], and [Reference Chen, Wang and Wu8–Reference Meng and Sharma10]. We note that the designed antennas are fed using a single short-ended CPW feed line. The following sections are dedicated to the design of the two antennas including all the simulation and measurement results.
II. DESIGN OF COMPACT DUAL-BAND ANTENNA WITH LARGE FREQUENCY RATIO
A) Antenna structure and design principles
All the designs of this paper are printed on the substrate F4B having a thickness of h = 1.964 mm, a relative permittivity of 2.55, and a loss tangent of 0.002. Figure 1 shows the structure of the single-feed dual-band antenna with large frequency ratio.
Fig. 1. Structure of the compact dual-band antenna with large frequency ratio (unit: mm); L 1 = 15.5, L 2 = 5.4, G = 0.1, L feed = 35.5. (a) Top view, (b) bottom view, (c) perspective view.
The top layer of the antenna contains two metallic patches shorted to the ground plane on one edge using metalized vias. The metalized vias have a diameter of 0.6 mm with a spacing of 1 mm between them. They act like a shorting wall, which reduces the length of the radiating patches to approximately a quarter wavelength instead of a half wavelength corresponding to an ordinary patch antenna without shoring edges. The width of the patches is arbitrarily chosen. In fact, it is found that the bigger the patch width the higher the antenna gain. The main criterion for our designs consists of setting the length of the patches L 1, and L 2 equal to a quarter wavelength. For this first antenna prototype, both patches have a width W = 20 mm and their lengths L 1, L 2 are set equal to a quarter wavelength according to the desired operating frequencies. As depicted in Fig. 2, at the resonant frequencies, the current flows from the top radiating patches to the ground plane through the metalized vias creating an open loop, which can be regarded as a magnetic dipole antenna [Reference He, Gong and Gao7, Reference Li, Lin, Chin, Liu and Cui14].
Fig. 2. Current distribution at the resonant frequencies of the antenna. (a) At 2.94 GHz, (b) at 7.5 GHz.
A single short-ended CPW line is used for the radio frequency (RF) feeding of the two bands. A good impedance value of 50-Ω is obtained by controlling the values of its length L feed and the gap G. As we shorted the radiating patches on one edge to the ground plane, the even resonant modes are suppressed so a large frequency ratio can be obtained [Reference Cabedo-Fabres, Antonino-Daviu, Valero-Nogueira and Bataller15]. We note that the disposition of the two patches is chosen in order to make simple the RF feeding structure. Indeed, the two patches are fed in series, which leads to a simple control of the impedance match by simply adjusting the length L feed. The designed antenna can also realize small frequency ratios as shown in the following parametric study.
B) Parametric study
The software used for the simulation is high frequency structure simulator based on finite elements method. The aim of this parametric study is to demonstrate that our dual-band antenna can be designed for large and small frequency ratios. We keep unchanged all the antenna parameters while we sweep the patch length L 2. Figure 3 illustrates the corresponding simulation results for the reflection coefficient |S 11|. The upper frequency band is changed according to the length L 2, which corresponds to a quarter wavelength. We note that the lower frequency band is not affected by changing the length L 2. This result demonstrates that the frequency ratio can be tuned according to the desired operating frequencies. After running a parametric study, an antenna prototype is fabricated in order to validate the simulation results.
Fig. 3. The simulated reflection coefficient |S 11| when L 2 is tuned.
C) Simulation and measurement results
An Agilent N5244A laboratory network analyzer is used to measure the antenna reflection coefficient |S 11|. Different antenna gains as well as radiation patterns are obtained in an anechoic chamber. Figure 4 illustrates the simulated and measured |S 11| of the antenna.
Fig. 4. Simulated and measured |S 11| of the dual-band with large frequency ratio.
The first measured operating frequency ƒ1 = 2.99 GHz corresponding to the lower band is nearly equal to the simulated value 2.94 GHz. The second measured operating frequency occurs at ƒ2 = 7.7 GHz with a good impedance match and 400 MHz of bandwidth evaluated at −10 dB. Above 6 GHz, the relative permittivity of the substrate used in this design (F4B) is less stable; this is why there is a frequency shifting between the simulated and measured values of ƒ2. A comparatively high frequency ratio ƒ2/ƒ1 = 2.58 is realized by this antenna. Figure 5 shows a sample of simulated and measured radiation patterns at ƒ1 and ƒ2. The maximum level of cross polarization obtained in E-plane is −18 dBi for both resonant frequencies. In H-plane, the level of cross polarization in the direction of maximum radiation is less than −22 dBi. The measured peak gains and efficiencies at ƒ1 and ƒ2 are respectively, 2.97 dBi–61% and 5.7 dBi–73%.
Fig. 5. Simulated and measured radiation patterns. (a) ƒ1 = 2.99 GHz in E-plane (φ = 0°), (b) ƒ2 = 7.7 GHz in E-plane (φ = 0°), (c) ƒ1 = 2.99 GHz in H-plane (φ = 90°), (d) ƒ2 = 7.7 GHz in H-plane (φ = 90°).
A comparative study of our work with some references is given in Table 1. Note that the lower resonant frequency is considered for computing the guided wavelength λg in the following table. According to Table 1 and compared with our design, the antenna presented in [Reference Chen, Wang and Wu8] has a large electrical size and a small frequency ratio of only 1.14. The design introduced in [Reference Meng and Sharma9] has a low gain value at the lower band and uses two PIN diodes to select only one operating frequency at one time. Although the antenna realizes 2.01 of frequency ratio, the use of PIN diodes will increase the complexity of the design once a DC bias circuit is introduced as well as decreasing the antenna efficiency. The antenna presented in [Reference Meng and Sharma10] has a very high thickness and uses a complex feeding system for the dual-band operation. We conclude that the designed antenna presents a large frequency ratio, an acceptable peak gain, and a compact size with simple RF feeding structure, which makes it a good candidate for wireless applications. In the following section, another dual-band antenna with improvement of the upper frequency bandwidth is further discussed.
Table 1. Comparison between our work and some references.
III. DESIGN OF COMPACT DUAL-BAND ANTENNA WITH BANDWIDTH ENHANCEMENT
A) Antenna structure and design principles
After designing a single-feed dual-band antenna with a large frequency ratio, the next step is to design another dual-band with improvement of the upper frequency bandwidth. The technique consists of modifying the RF feed line structure in order to create another resonant frequency. The antenna structure is given in Fig. 6.
Fig. 6. Structure of the compact dual-band with bandwidth enhancement (unit: mm); L 1 = 15.6, L 2 = 6.7, L g = 34.1, W g = 0.4. (a) Top view, (b) bottom view.
The radiating patches on the top side of the antenna have a width W = 26 mm. The lengths of the patches are chosen to excite two operating frequencies 2.84 and 6.19 GHz. The geometrical dimensions of the feed line denoted by W g and L g are adjusted carefully to get the desired resonant frequency in the upper band. By combining the resonant frequency obtained through the radiating patch L 2 and the one created by modifying the RF feed line structure, we get a wide upper bandwidth. A parametric study showing the effects of W g and L g on the resonant frequency of the upper band is carried out. We note that the impedance match is controlled by adjusting the dimensions of the feed line. The antenna overall size is 40 ×40 × 1.964 mm3.
B) Parametric study
This parametric study aims to investigate the influence of the feed line geometrical parameters denoted by W g and L g on the antenna resonant frequencies. First, all the antenna parameters are kept unchanged and only W g is changed.
According to Fig. 7, when the feed line is not modified (kept the same as the structure shown in Fig. 1) with W g = 2.6 mm, the antenna presents an upper impedance bandwidth of only 130 MHz (2.06%) evaluated at −10 dB of |S 11| and centered at 6.3 GHz. By decreasing the value of W g, another resonant frequency appears just near the operating frequency excited by the patch L 2. The largest bandwidth is obtained when W g = 0.4 mm. Then, the length L g is adjusted for a good impedance matching as shown in Fig. 8.
Fig. 7. The simulated reflection coefficient |S 11| when W g is tuned.
Fig. 8. The simulated reflection coefficient |S 11| when L g is tuned.
In fact to realize a wide upper bandwidth, two resonant frequencies are carefully adjusted and combined together. The first resonant frequency is controlled and excited by the patch L 2 while the second resonant frequency is excited by the feeding line, which creates an additional resonant mode. By combining two resonant modes, one created by the patch L 2 and the second one created by the feeding line, a wide bandwidth with better impedance match can be obtained. To demonstrate that there are two resonant frequencies in the upper band, the input impedance Z 11 is displayed. The second resonant frequency can be controlled and adjusted by tuning the value of W g. According to Fig. 9, two resonant frequencies are presented, the first one occurs at 6.2 GHz corresponding to the patch L 2; It remains unchanged when W g is tuned. The second resonant frequency increases by increasing the width W g. We note also from Fig. 9 that by reducing the line width W g the two resonant frequencies get closer and can be combined to realize a wide frequency bandwidth.
Fig. 9. The simulated input impedance when W g is tuned.
C) Simulation and measurement results
An antenna prototype is fabricated to validate the proposed approach. The simulated and measured reflection coefficient |S 11| of the antenna as well as the radiation patterns are carried out. Figure 10 shows the simulated and measured |S 11|. The measured result of the |S 11| reveals two frequency bands, the lower band is centered at 2.93 GHz and the upper band ranges from 6.2 to 6.8 GHz (9.23% at the central frequency 6.5 GHz) evaluated at −10 dB. An important bandwidth enhancement from 2.06 to 9.23% is obtained by modifying the feed line structure of this antenna. Some samples of simulated and measured radiation patterns are shown in Fig. 11. The minimum measured peak gain is 2.95 dBi for the operating frequency 2.93 GHz and goes from 5.3 to 6.1 dBi for the band 6.2–6.8 GHz. The antenna measured efficiencies take values in the interval [61, 74%].
Fig. 10. Simulated and measured |S 11| of the dual-band with bandwidth enhancement.
Fig. 11. Simulated and measured radiation patterns. (a) ƒ1 = 2.93 GHz in E-plane (φ = 0°), (b) ƒ2 = 6.5 GHz in E-plane (φ = 0°), (c) ƒ1 = 2.93 GHz in H-plane (φ = 90°), (d) ƒ2 = 6.5 GHz in H-plane (φ = 90°).
To demonstrate the importance of this design, a comparative study with other references is displayed in Table 2. According to Table 2 and compared with [Reference Zhu, Guo and Wu3], our design uses a simple CPW RF feeding line, which reduces the fabrication complexity. Also the bandwidths for the lower and upper frequency bands are much higher than the values realized by the design of [Reference Zhu, Guo and Wu3]. In [Reference Chen, Wang and Wu8], the antenna realizes a value of 5.5% of bandwidth at the lower band. However, the RF microstrip feed line have to be properly located; otherwise the antenna will resonate only at a single frequency. In our design, the RF feeding line is very easy to adjust by a simple control of its length and width. In [Reference Meng and Sharma9], beside the low values of the bandwidths, the antenna gain is found to be very poor at the lower frequency band compared with the value obtained in our design. We deduce that our antenna has better performance including compact size and wide upper impedance bandwidth with a very simple CPW RF feed line.
Table 2. Comparison between our work and other papers.
IV. CONCLUSION
In this paper, two antennas are presented and discussed. The first one is a single-feed dual-band antenna having a large frequency ratio of 2.58. The second antenna has the particularity of realizing a comparatively wide impedance bandwidth of 9.23% at the central frequency 6.5 GHz. A simple RF single-feed technique leading to ease of fabrication and compact size is adopted. The two designs have an acceptable measured gains and efficiencies with low profile. Therefore, they can find their use in wireless communication applications.
ACKNOWLEDGEMENTS
This work was supported in part by NSFC (No. 61571084), in part by NCET (No. NCET-13-0095), in part by the FRF for CU (No. ZYGX2014J016), and in part by the SRF for ROCS, SEM.
Abdelheq Boukarkar was born in Algeria on November 11, 1984. He received the B.S. degree in 2009 and the M.S. degree in 2015 from the University of Electronic Science and Technology of China (UESTC) and he is working toward the Ph.D. degree in Electronic Engineering. His research interests include antenna design and reconfigurable RF/Microwave circuits.
Xian Qi Lin was born in Zhejiang Province, China, on July 9, 1980. He received the B.S. degree in Electronic Engineering from UESTC (University of Electronic Science and Technology of China), in 2003, and Ph.D. degree in Electromagnetic and Microwave Technology from Southeast University, Nanjing, China, in 2008. He joined the Department of Microwave Engineering at UESTC in August 2008, and has become an Associate Professor and a doctoral supervisor since July 2009 and December 2011, respectively. From September 2011 to September 2012, he was a post-doc researcher in the Department of Electromagnetic Engineering at Royal Institute of Technology (KTH), Sweden. He has authored over 10 patents, over 40 scientific journal papers, and has presented over 20 conference papers. His research interests include microwave/millimeterwave circuits, and antennas. Dr. Lin is a member of IEEE and a reviewer of many well-known journals such as IEEE- MTT/AP/MWCL/AWPL, JEMWA/PIER and EL.
Yuan Jiang was born in Jiangsu China. He received the B.S. degree in 2006 and is working toward the Ph.D. degree in Electronic Engineering from the University of Electronic Science and Technology of China (UESTC). His research interests include reconfigurable RF/Microwave circuits and antennas.
